Fast-response current limiting

ABSTRACT

An amplifier ( 10″ ) has a first amplifier stage ( 14 ) for producing a control current (I X ) in response to an input voltage. A second amplifier stage ( 16 ) has first ( 46 ) and second ( 38 ) transistors. The first transistor ( 46 ) is coupled to receive the control current (I X ) and is operable to produce a control voltage. The second transistor ( 38 ) is coupled to receive the control voltage and operable to produce an output current. A nonlinear resistive element ( 50 ) is coupled to the first transistor ( 46 ) to add a nonlinear function of the control current (I X ) to the control voltage. The nonlinear resistive element ( 50 ) may include a third transistor connected between the first transistor ( 46 ) and a reference potential, operable to receive the control current (I X ) and to generate the nonlinear function thereof.

BACKGROUND OF INVENTION

1. Field of Invention

This invention relates to improvements in electrical circuits, and moreparticularly to circuits and methods for fast response current limiting.

2. Relevant Background

A prior-art two-stage amplifier 10 is shown in FIG. 1 for driving anoutput circuit 12, having a load, R_(L), 18 and an nMOS output powertransistor, M_(POWER), 20. The circuit 10 may be, for example, anintegrated circuit, with the circuit 12 provided either as a partthereof, or externally connected thereto.

The amplifier 10 has two stages 14 and 16. The first amplifier stage 14has two PNP transistors Q₁, 22, and Q₂, 24, which have an emitter arearatio of 1X/NX. The transistors 22 and 24 have respective associatednMOS current mirror transistors M₂, 26, and M₃, 28. The transistors 26and 28 mirror the current in nMOS transistor M₁, 30, which is connectedto current source 32 that provides a current I_(B) thereto. If the nMOStransistors 26 and 28 are designed to conduct equal currents, aninput-referred voltage offset V_(PTAT) will occur between the PNPtransistors 22 and 24 due to their size differences and resultingdifferent current densities therein. Assuming transistors 22 and 24operate in low-level injection, the magnitude of the voltage offsetequalsV _(PTAT) =V _(r)1n(N)  [1

-   -   where V_(T) is the thermal voltage (V_(T)=kT/q, where k is        Boltzmann's constant, T is the absolute temperature, and q is        the charge on the electron), and where N is the emitter ratio as        described above).

Since the circuit 10 is preferably fabricated as a single integratedcircuit, the resistor R_(AI) across which the input voltage is developedis preferably constructed of aluminum metallization, which has atemperature coefficient of about 3000 ppm/° C. The temperaturecoefficient of the resistor then matches (at least approximately) thetemperature coefficient of the voltage V_(PTAT). Therefore the currentlimit I-_(lim) generated by circuit 10 is largely independent oftemperature. This current limit equals $\begin{matrix}{I_{\lim} = \frac{V_{T}{\ln(N)}}{R_{AI}}} & \lbrack 2\rbrack\end{matrix}$

In response to the input voltage across R_(AI), the first amplifierstage 14 generates a current I_(X) that is sunk into MOS transistor M₄of the second amplifier stage 16. The magnitude of current I_(X) equals:$\begin{matrix}{I_{X} = {I_{S}\quad{\exp\left( \frac{V_{IN} - V_{PTAT}}{V_{T}} \right)}}} & \lbrack 3\rbrack\end{matrix}$

-   -   where I_(s) is the saturation current of Q₁, and V_(IN) equals        the differential voltage appearing across resistor R_(AI). The        current I_(X) is referred to herein as a “control current”.

The second amplifier stage 16 includes a first nMOS transistor M₄, 46,connected to receive the control current I_(X), and a second nMOStransistor M₅, 38, connected to a current source 40, which provides acurrent I_(CP).

The first transistor 46 amplifies the control current I_(X) to develop acontrol voltage, which is applied to the gate of the second transistor38. The second transistor 38 generates an output current, which, in theembodiment illustrated controls the voltage at the gate of the outputpower transistor 20.

The frequency response of the amplifier of circuit 10 has two importantpoles. The first is an internal pole caused by capacitance C_(X) actingagainst the resistance at node 36. The second is a gate pole caused bythe capacitance C_(g) acting against the resistance at node 43. In orderto maintain adequate stability, the gain of the circuit must drop belowunity before the phase margin drops below about 30°. This requireseither that one of the poles be pushed back to a very low frequency(dominant-pole compensation) or that the gain of the circuit beartificially reduced.

Dominant-pole compensation is greatly complicated by the movement of thegate pole due to variations in effective gate capacitance C_(g) withload resistance R_(L). If R_(L) is shorted, then C_(g) includes a largecontribution from the gate-to-source capacitance C_(gs) of the outputpower transistor 20. Larger values of R_(L) decrease the contribution ofC_(gs) to C_(g). Dominant-pole compensation can still be achieved eitherby adding a large capacitance to node 43, or by connecting a Millercapacitance around transistor 38, but both solutions have theundesirable property of slowing the transient response of the amplifier.

The addition of transistor 46 greatly reduces the resistance at node 36.This has two beneficial effects. First, it reduces the loop gain, andsecond, it pushes the internal pole out to a higher frequency,effectively forcing the gate pole to become the dominant pole of thesystem. The addition of transistor 46 therefore compensates theamplifier without requiring the addition of any extraneous capacitance.The offset introduced by current I_(X) can be compensated by drawing anequal current from node 45.

The circuit of FIG. 1 responds relatively rapidly to large inputsignals, such as those generated by hot-shorting the load R_(L). Thecurrent available to slew the gate capacitance is the current intransistor 38, which equals $\begin{matrix}{I_{M5} = {\frac{\left( {W/L} \right)_{5}}{\left( {W/L} \right)_{4}}I_{X}}} & \lbrack 4\rbrack\end{matrix}$

-   -   where (W/L)₅ and (W/L)₄ are respectively the width to length        ratios of the nMOS transistors M₅, 38, and M₄, 46. As equation        [3] indicates, this current is exponentially dependent upon the        magnitude of the input voltage. This equation does not consider        the terminal resistances of transistors 22 and 24, nor their        finite betas. These factors will ultimately limit the current        I_(X), and through it, the response time of circuit 10.

SUMMARY OF INVENTION

In light of the above, the invention, in accordance with a broad aspectthereof, presents an amplifier. The amplifier has a first amplifierstage for producing a control current. in response to an input voltage.A second amplifier stage has first and second transistors. The firsttransistor is coupled to receive the control current and is operable toproduce a control voltage. The second transistor is coupled to receivethe control voltage and is operable to produce an output current. Anonlinear resistive element is coupled to the first transistor to add anonlinear function of the control current to the control voltage. In oneembodiment, the nonlinear resistive includes a third transistorconnected between the first transistor and a reference potential, thethird transistor operable to receive the control current and to generatethe nonlinear function of the control current.

In accordance with another broad aspect of the invention, a circuit ispresented that has an amplifier for producing a control current inresponse to an input voltage and a control voltage in response to thecontrol current. A circuit is provided for producing an output currentin response to the control voltage. A nonlinear resistive element isalso provided for adding voltage that is a nonlinear function of thecontrol current to the control voltage. An output stage is driven inresponse to the output current.

In accordance with still another broad aspect of the invention, a methodis presented for controlling an output current of a circuit. The methodincludes producing a control current in response to an input voltage anda control voltage in response to the control current, producing anoutput current in response to the control voltage, providing a nonlinearresistive element for adding voltage that is a nonlinear function of thecontrol current to the control voltage, and driving an output stage inresponse to the output current.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a two-stage amplifier circuit, according to the prior art.

FIG. 2 is a two-stage amplifier circuit showing an example of anincorporation therewith of a nonlinear resistive element, according to apreferred embodiment of the invention.

FIG. 3 is a two-stage amplifier circuit additionally showing circuitryfor sinking additional currents, according to another preferredembodiment of the invention.

FIG. 4 is a two-stage amplifier circuit further showing circuitry forreducing capacitive effects of at least some circuit elements, accordingto still another preferred embodiment of the invention.

In the various figures of the drawing, like reference numerals are usedto denote like or similar parts.

DETAILED DESCRIPTION

With reference now additionally to the circuit 10′ of FIG. 2, a thirdnMOS transistor M₆, 50, is shown connected between the source of thefirst nMOS transistor 46 and ground. The third transistor 50 may be annMOS device, as in the embodiment shown. Those skilled in the art willrecognize that the circuit 10′ may be constructed with transistors ofdifferent types, (i.e., lateral, vertical, bipolar, MOSFET, and on) andconductivities (i.e., PNP, NPN, nMOS, pMOS, and so on), and that othercircuit components (not shown) for collateral purposes may also beemployed within the circuit 10′.

First transistor 46, although necessary in order to ensure stability,slows the transient response of circuit 10′. The first transistor 46draws off current that could otherwise transiently charge capacitance42. Furthermore, since the current provided by PNP transistor 22 islimited by beta and terminal resistances, the first transistor 46effectively clamps the voltage swing seen at the node 36, and thuslimits the maximum current that the second transistor 38 can sink. Theslower transient performance increases the time required to turn off theoutput transistor 20 in the event of a severe overcurrent condition,such as a short circuit. However, these and other issues are addressedby the addition of the third transistor 50.

The third transistor 50 is sized to provide a much smallerdrain-to-source resistance than the first transistor 46 when the circuit10′ operates at or near equilibrium (V_(IN)≈V_(PTAT)). In the embodimentshown, this entails sizing the third transistor 50 so that it operatesin the triode (or linear) region when the circuit is at or nearequilibrium, and further constructing the third transistor 50 to have amuch larger width-to-length ratio than the first transistor 46.

The drain-to-source resistance of the third transistor 50 willdramatically increase when the current flowing through it exceeds athreshold value. In the embodiment shown, this increase in resistancecorresponds to the transition from the triode region to the saturationregion. Thus, the third transistor 50 is sized so that this transitionoccurs only when the circuit is perturbed from its equilibrium condition(V_(IN)>>V_(PTAT)).

In or near equilibrium, the drain-to-source resistance of thirdtransistor 50 is much smaller than that of the first transistor 46.Thus, circuit 10′ operates essentially in the manner as theabove-described circuit 10. However, if circuit 10′ is perturbed fromequilibrium, for example by a short-circuiting of the load resistanceR_(L), then the current I_(X) will increase to the point that the thirdtransistor 50 enters saturation. At this point, the current flow throughthe first transistor 46 is effectively choked off by the largedrain-to-source resistance of the third transistor 50. Any increase incurrent I_(X) above and beyond that required to saturate the thirdtransistor 50 will then go to charge capacitance 42 and slew node 36.Furthermore, the voltage of node 36 will no longer be limited by thecurrent that the PNP transistor 22 can deliver. Therefore circuit 10′will slew substantially faster than prior-art circuit 10.

When the third transistor 50 saturates, the pole created by parasiticcapacitor 42 moves in to lower frequencies. If the circuit wereoperating at equilibrium, this would erode the phase margin and couldpotentially cause the circuit to become unstable. However, thirdtransistor 50 only saturates when circuit 10′ is far from equilibrium.As the circuit approaches equilibrium, the third transistor 50 dropsback into triode mode, and the pole created by parasitic capacitor 42moves out to higher frequencies. Therefore circuit 10′ exhibits stableoperation at equilibrium in combination with rapid slewing when far fromequilibrium.

More generally, the third transistor 50 may be viewed as a nonlinearresistive element that adds a nonlinear function of the control current,I_(X), to the control voltage that is generated by the first transistor46. This nonlinear function is only weakly dependent upon the controlcurrent up to a certain threshold, beyond which the control voltageincreases very rapidly as a function of the control current. Thisthreshold is selected so as to lie well above the control current,I_(X), expected to flow under equilibrium conditions. It should beunderstood that the third transistor 50 can be replaced with any circuitelement that operates in a manner similar to that described above. Thisresults in the advantages described above, and, more particularly,extends the use of the circuit of FIG. 1 to enable it to be used withdiscrete power transistors which operate at much larger voltages.

It should be emphasized that although the third transistor 50 is shownand described above in the context of a MOS device, and moreparticularly, and nMOS device, it may be a pMOS device, a bipolartransistor, or other appropriate device. If a bipolar transistor is usedfor the third transistor 50, it would be biased such that at or nearequilibrium, the transistor would operate in a saturation mode, and awayfrom equilibrium, the transistor would operate in a forward-active mode.The bipolar transistor also can be either a PNP or NPN device, dependingupon the particular circuit construction employed.

It should be noted that the output power transistor 20, particularly ifit is externally provided, should have a low on-resistance in order toprevent excessive conduction losses. However, during a hot-short event,this low on-resistance may allow extremely large currents to flow. Themagnitude of these currents, coupled with the large voltages presentacross the transistor, can produce extreme levels of power dissipation,which in some cases may be in the kilowatt range. In such cases, theoutput power transistor 20 needs to be turned off very quickly,typically within a microsecond or two, in order to prevent itsdestruction. Moreover, a large external output power transistor may havea correspondingly large gate capacitance, which makes it even moredifficult to turn off quickly. Thus, an amplifier circuit that can sinklarge currents and slew rapidly is of great practical significance forcurrent limiting applications.

The addition of the third transistor 50 improves the slew rate responseof the circuit 10′ and (in most cases) increases the maximum currentthat the second transistor 38 can sink. However, the benefits of thethird transistor 50 are limited by certain practicalities of circuitdesign, most notably the sizing requirements for transistor 38.

If additional sinking current is desired, additional circuitry may beadded, as shown in the circuit 10″ of FIG. 3, to which reference is nowadditionally made. Circuit 10″ includes a booster circuit comprising afourth nMOS transistor M₇, 52. Although an nMOS device is shown, thoseskilled in the art will appreciate that other types and conductivitiesof devices may be used, depending upon the particular construction ofthe circuit 10″. The fourth transistor 52 may be a large device that cansink a correspondingly large current. When circuit 10″ operates at ornear equilibrium, the voltage developed across third transistor 50 isinsufficient to bias the fourth transistor 52 into conduction.Furthermore, the large gate capacitance of the fourth transistor 52 isshunted to ground through the relatively low drain-to-source resistanceof the third transistor 50, effectively suppressing any pole or zerothat this gate capacitance might otherwise have generated. Therefore,while at or near equilibrium, circuit 10″ acts in much the same way ascircuit 10.

When the third transistor 50 is biased into saturation, the potential onnode 53 rises, turning on the fourth transistor 52. The fourthtransistor 52 provides additional sinking current to help pull the gateof the output power transistor 20 to ground. This action is in additionto the action of the second transistor 38 of the second amplifier stage16, which also serves to pull down the gate of the output powertransistor 20 to ground.

In the operation of the circuit 10″, when the third transistor 50saturates, current I_(X) charges the gate capacitance of the fourthtransistor 52. If the fourth transistor 52 is very large, its gatecapacitance may slow the slew of the circuit, thereby slowing theoverall response time. In such cases, the circuit embodiment 10′″ ofFIG. 4, to which reference is now additionally made, may be used. In thecircuit 10′″, the gate of the fourth transistor M₇, 52, is connected toa node V_(z), 59, between fifth and sixth nMOS transistors M₈, 54, andM₉, 56, connected between the supply voltage 48 and ground. Fourth,fifth, and sixth transistors 52, 54 and 56 together form a boostercircuit for the circuit embodiment 10′″.

The gate of the fifth transistor 54 is connected to the gate of thefirst transistor 38 of the second amplifier stage 16, and the gate ofthe sixth transistor 56 is connected to the gate of the third transistor50. The drain of the fifth transistor 54 is connected to the supplyvoltage 48, and the sixth transistor 56 is connected between the sourceof the fifth transistor 54 and ground. In the embodiment shown,transistors 54 and 56 may be nMOS devices, as shown; however, as above,it will be appreciated by those skilled in the art, other types andconductivities of devices may be used, depending upon the particularconstruction of the circuit 10′″.

In the circuit 10′″, the fifth transistor 54 acts as a source follower,which is biased into conduction by the sixth transistor 56. Theinsertion of a source follower provides a much larger current to chargethe gate capacitance of the fourth transistor 52, without greatlyincreasing the capacitance seen at node 36. What capacitance is seen atnode 36 can be minimized by making the fifth transistor 54 relativelywide and narrow, thus maximizing its transconductance for a given gatecapacitance. This is allowable since the fifth transistor 54 does notneed to accurately match any other transistor in the circuit.

Circuits 10′, 10″ and 10′″ represent a progressive development of asingle concept. All three circuits contain a nonlinear resistive element(50) that allows a rapid increase in the control voltage (at node 36)when the circuit is driven from equilibrium. All three circuits achievea faster rate of slew on the inter-stage node (node 36) through use ofthe nonlinear resistive element, and all three achieve highersecond-stage sink currents, although they differ in their means towardsthis end. Circuit 10′ relies merely upon a high voltage at node 36 fullyenhancing transistor 38. Circuit 10″ supplements transistor 38 with abooster circuit that includes a transistor 52 which conducts only afterthe nonlinear resistive element has transitioned from its low-resistanceregion to its high-resistance region. Circuit 10′″ uses a boostercircuit that includes a source follower stage comprising transistors 54and 56 to enable a much larger output sink device (transistor 52)without excessive increase of the capacitance on node 36.

It should be noted that although the circuits of FIGS. 2, 3, and 4 areshown in the context of the transconductance and load circuits 14 and 12of FIG. 1, the transconductance and load circuits shown are examplesonly, and that various other circuits and circuit arrangements may beused in place thereof within the scope of the invention. Moreover, otherpermutations may be used; as suggested above, for example, the circuitsmay be easily implemented entirely in bipolar devices or entirely in MOSdevices. Another easily implemented permutation may be the inversion ofthe power supply, wherein a negative power supply may be used referencedto ground or to a positive potential, with appropriate polarity changesof the devices of the circuit.

Although the invention has been described and illustrated with a certaindegree of particularity, it is understood that the present disclosurehas been made only by way of example, and that numerous changes in thecombination and arrangement of parts can be resorted to by those skilledin the art without departing from the spirit and scope of the invention,as hereinafter claimed.

1-22. (canceled)
 23. A method for controlling an output current of acircuit, comprising: producing a control current in response to an inputvoltage and a control voltage in response to said control current;producing an output current in response to said control voltage;providing a nonlinear resistive element for adding voltage that is anonlinear function of said control current to said control voltage; anddriving an output stage in response to said output current.
 24. Themethod of claim 23 further comprising operating said nonlinear resistiveelement in a low dynamic resistance mode when said control current isbelow a predetermined level and in a high dynamic resistance mode whensaid control current is above said predetermined level.
 25. The methodof claim 23 further comprising providing a parallel path for said outputcurrent when said control current exceeds a predetermined value.
 26. Themethod of claim 25 further comprising isolating a capacitance of anelement of said parallel path.